Single stage switching power converter with improved primary only feedback

ABSTRACT

A switching power converter is provided that extrapolates from a reference voltage during dead periods of a rectified input voltage as determined from a comparison of an Isense voltage to a current threshold.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.15/423,335, filed Feb. 2, 2017, which in turn is a continuation ofInternational Application No. PCT/CN2015/083608, filed Jul. 8, 2015.

TECHNICAL FIELD

This application relates to switching power converters, and moreparticularly to a switching power converter with improved primaryfeedback.

BACKGROUND

Single-stage AC-DC power conversion is a low cost and thus popular powersupply topology. In a single-stage AC-DC power converter, the AC linevoltage is rectified to produce a rectified input voltage that cyclesfrom approximately zero volts to the peak line voltage (e.g., 1.414*120V in the US) at twice the AC frequency. Single-stage AC-DC switchingpower converters include a power switch that cycles multiple timesduring each cycle of the rectified input voltage. Each time the powerswitch cycles, a pulse of power is delivered to the load. During thebulk of each rectified input voltage cycle, the rectified input voltagelevel is relatively high such that a significant amount of power isdelivered to the load with every cycle of the power switch. But duringthe “dead” period of each rectified input voltage cycle in which therectified input voltage weakens and drops to 0 V, the power deliverywith each cycle of the power switch is relatively weak.

This weak power delivery during the dead period between the rectifiedinput voltage peaks complicates the output voltage regulation inindirect control topologies that use a sense voltage from an auxiliarywinding such as practiced in single-stage flyback power converters usingprimary-only feedback. Flyback converters are commonly used as theswitching power converter in a single-stage AC-DC architecture. But theisolation between the output voltage on the secondary side of thetransformer and the primary side of the transformer in a flybackconverter complicates its regulation. The output voltage may be sensedusing optoisolators but that raises costs and control complication. Incontrast, primary-only feedback control techniques determine the outputvoltage by sampling the reflected voltage on an auxiliary winding at thetransformer reset time. When the rectified input voltage is relativelyhigh, the pulse of energy delivered to the load at each cycle of thepower switch is relatively robust such that there is a linearrelationship between the reflected voltage and the output voltage. Butthis linear relationship becomes muddied for the “runt” pulses deliveredto the load during the dead period between consecutive rectified inputvoltage peaks. The primary-only feedback loop thus operates witherroneous output voltage information during the dead periods due to thebreakdown in the linear relationship between the reflected voltage andthe output voltage for the runt pulses.

This degradation for primary-only feedback control systems isproblematic in that a single-stage AC-DC switching power converter maybe designed for world-wide use to lower costs and take advantage of massproduction efficiencies. But the power line cycling varies across theworld depending upon the vagaries of a particular country's electricalpower providers. For example, the United States operates with a 60 Hz120 V (RMS) AC main whereas many other countries such as in Europeoperate with a 50 Hz 230 V (RMS) AC main. The control loop in asingle-stage AC-DC switching power converter designed for world-wideusage must accommodate these diverse inputs while still keeping theiroutput power to the load within the desired regulation. But suchregulation is weakened due to the dead periods of the rectified inputvoltage cycles for feedback systems that use a sense voltage from anauxiliary winding instead of sampling the output voltage directly suchas through optoisolators.

Accordingly, there is a need in the art for improved primary-onlyregulation of single-stage AC-DC switching power converters.

SUMMARY

A single-stage switching power converter is provided that includes atracking circuit that adjusts a reference voltage to track an outputvoltage using a sense voltage from an auxiliary winding. Thesingle-stage switching power converter also includes a comparator forcomparing an on-time current produced from a power switch on-time to acurrent threshold to determine when the sense voltage is trustable andwhen the sense voltage is non-trustable. The single-stage switchingpower converter includes an extrapolator that determines in each cycleof a rectified input voltage a first trustable value for the referencevoltage and a last trustable value for the reference voltage responsiveto when the sense voltage is trustable and non-trustable as determinedby the comparator. The extrapolator extrapolates from the firsttrustable value and the last trustable value to provide an extrapolatedreference voltage while the sense voltage is non-trustable thatsubstantially equals the output voltage The resulting extrapolation isquite advantageous because the single-stage switching power convertermay control its power switch while the sense voltage is non-trustableusing the extrapolated reference voltage. Since this extrapolatedreference voltage substantially equals the output voltage while thesense voltage is non-trustable, the output power regulation isconsiderably improved as compared to a conventional techniques.

These advantageous features may be better appreciated through aconsideration of the detailed description below.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A illustrates a linear and a curvilinear extrapolation of thereference voltage during the dead period between two cycles of therectified input voltage in accordance with an embodiment of thedisclosure.

FIG. 1B illustrates the reference voltage for two cycles of therectified input voltage without any extrapolation.

FIG. 1C illustrates a constant-level extrapolation of the referencevoltage during the dead period between two cycles of the rectified inputvoltage in accordance with an embodiment of the disclosure.

FIG. 2 illustrates a flyback converter including a controller configuredto practice a reference voltage extrapolation in accordance with anembodiment of the disclosure.

FIG. 3 illustrates the Vsense and Isense voltage waveforms for both aVsense OK period of the rectified input voltage and for a dead period ofthe rectified input voltage in accordance with an embodiment of thedisclosure.

FIG. 4 is a block diagram of the controller of FIG. 2 in accordance withan embodiment of the disclosure.

FIG. 5A illustrates a Vref extrapolation during a dead period in whichthe load is increasing its power consumption.

FIG. 5B illustrates a Vref extrapolation during a dead period in whichthe load is decreasing its power consumption.

FIG. 6 illustrates a generic power converter including a controllerconfigured to extrapolate from the reference voltage during dead periodsof the rectified input voltage in accordance with an embodiment of thedisclosure.

Embodiments of the present disclosure and their advantages are bestunderstood by referring to the detailed description that follows. Itshould be appreciated that like reference numerals are used to identifylike elements illustrated in one or more of the figures.

DETAILED DESCRIPTION

To address the need for improved regulation in single-stage AC-DCswitching power converters with primary-only output voltage regulation,the current through the power switch during the power switch cycles iscompared to a threshold to distinguish between relatively-high-powerpower pulses delivered to the load versus the delivery of low-powerpower pulses. This improved regulation may be applied to any suitableswitching power converter architecture including flyback, buck, andboost switching power converter architectures. The following discussionwill be directed to a flyback converter topology without loss ofgenerality in that the same principles may be applied to othersingle-stage switching power converter architectures that use a sensevoltage obtained from an auxiliary winding such as performed inprimary-only feedback systems. For those power pulses that satisfy thethreshold, the output voltage regulation occurs in a conventionalfashion in that the output voltage is represented by a reference voltageVref that is compared to the primary winding reflected voltage (Vsense).As the primary winding reflected voltage changes due to the changingamounts of power delivered to the load in each cycle of the powerswitch, the reflected voltage changes to track the output voltage (Vout)applied to the load. Such tracking is conventional and may be performedusing a gap judgment as explained further herein.

But as the rectified input voltage cycles into a dead period, the powerpulses delivered to the load become smaller and smaller. The resultingreflected voltages on the primary winding during the off period in eachpower switch cycle will then no longer accurately reflect the outputvoltage Vout. Such faulty reflected voltage pulses are denoted herein as“runt” pulses. The peak current through the primary winding during eachon period of the power switch cycle will also fall as the rectifiedinput voltage falls such that the peak current will fall below thecurrent threshold at a time denoted herein as the “last trustablepoint.” The corresponding reference voltage at this time may thus bedenoted as the “last trustable voltage reference.” As the rectifiedinput voltage begins again to cycle high, the peak current will increaseuntil it exceeds the current threshold at a time denoted herein as the“first trustable point.” The corresponding reference voltage at thefirst trustable point may thus be denoted as the “first trustablereference voltage.”

In any given cycle of the rectified input voltage, there will thus be acurrent first trustable reference voltage and a current last trustablereference voltage. These two values may be used to extrapolate what thereference voltage should be from the time of the current last trustablereference voltage in a given rectified input voltage cycle until asubsequent first trustable reference voltage occurs in a subsequentrectified input voltage cycle. This extrapolation may be betterappreciated with regard to FIG. 1A, which shows a current cycle for therectified input voltage Vin followed by a subsequent cycle of therectified input voltage Vin. In addition, the reference voltage Vrefresulting from these two cycles of the rectified input voltage is alsoshown in FIG. 1A. A current threshold for the primary winding current(not illustrated) is used to detect when the current first trustableVref value occurs in the current cycle of the rectified input voltageVin. At this point, the reference voltage Vref is said to “toggle” inthat it is changed from power-switch-cycle-to-power-switch-cycle in aconventional fashion as discussed earlier. But as the primary windingcurrent again drops below the current threshold, the current lasttrustable Vref value is detected in the current cycle of the rectifiedinput voltage Vin. Until the subsequent first trustable Vref value isdetected in the subsequent cycle of the rectified input voltage Vin, thesensed voltage (not illustrated) is no longer trustable as discussedearlier. But it may be assumed during this dead period of the rectifiedinput voltage that the reference voltage should continue following itsregular sinusoidal pattern that is induced from the sinusoidal profileof the rectified input voltage. During the dead period, it may thus beassumed that the subsequent first trustable value for Vref will equalthe current first trustable value for Vref. Moreover, the duration ofthe dead period may be assumed to be the same from cycle-to-cycle of therectified input voltage Vin. Given these reasonable assumptions, alinear extrapolation for Vref for each power switch cycle in the deadperiod equals: Vref (linear extrapolated)=current last trustable valuefor Vref−((current first trustable value for Vref−current last trustablevalue for Vref)*(current expired fraction of the dead period). The“current expired fraction of the dead period” represents the fraction ofthe dead period that has expired at a given time within the dead periodfor the current cycle of the rectified input voltage Vin. For example,for a power switch cycle occurring in the exact middle of the deadperiod, the current expired fraction would be 0.5. Similarly, for apower switch cycle at the end of the dead period, the current expiredfraction would be 1, and so on. Rather than digitally calculate thelinear extrapolation, analog methods may be used. For example, a rampgenerator may be configured to linearly ramp an extrapolated Vref overthe dead period, wherein the linear ramp starts from the current lasttrustable value for Vref and ends at the current first trustable valuefor Vref. Similarly, both digital and analog techniques may be used toprovide a curvilinear extrapolation of Vref.

The resulting linear extrapolation for Vref during the dead period isrepresented by a line 100 in FIG. 1A. Alternatively, a curvilinearfitting may be used to extrapolate Vref during the dead period asrepresented by a line 105. Regardless of whether Vref is linearly orcurvilinearly extrapolated between the first and last trustable valuesfor Vref during the dead period, note how advantageous the resultingregulation is over a conventional method such as shown in FIG. 1B. Inthis conventional method, Vref continues to track the sensed voltageeven during the dead period. As a result, Vref drops substantially tozero during the dead period since Vref is being toggled with regard to anon-trustable primary winding reflected voltage during the dead period.The resulting conventional regulation during the dead period is thusbased upon a Vref that is no longer representative of the output voltageduring the dead period. In sharp contrast, the use of the extrapolatedVref during the dead period as discussed with regard to FIG. 1A providesa much more accurate representation of the output voltage and thusprovides a tighter regulation of the output voltage (or output current).

Rather than use an extrapolated reference voltage, the current lasttrustable reference voltage may be detected using the current thresholdon the primary winding current and then maintained constant during thedead period as shown in FIG. 1C for a flat line 110 until the subsequentfirst trustable reference voltage is detected using the currentthreshold on the primary winding current. At that point, theconventional toggling of the reference voltage would commence startingfrom the first trustable reference voltage value. Although such aconstant Vref is more accurate than the prior art method discussed withregard to FIG. 1B, a comparison of flat line 110 with either linearextrapolated Vref 100 or curvilinear extrapolated Vref 105 of FIG. 1Ashows that a linear (or curvilinear) extrapolation of the referencevoltage during the dead period provides much better regulation of theoutput voltage during the dead period.

An example single-stage flyback converter 200 is shown in FIG. 2. Acontroller integrated circuit 205 for flyback converter controls theswitching of a power switch such as an NMOS transistor Q1. Inalternative embodiments, the power switch may be implemented using abipolar junction transistor. Controller 205 controls a gate voltage forpower switch transistor Q1 through an output pin 5. When switched on,power switch transistor Q1 allows a primary current to flow through aprimary winding 210 of a transformer 215 responsive to the rectifiedinput voltage Vin. A rectifier 220 including a diode bridge and acapacitor C1 rectifies the alternating line voltage carried on an ACmain 224 to provide the rectified input voltage Vin. The rectified inputvoltage Vin thus still retains a pronounced sinusoidal profile acrosseach half of an AC main cycle.

For each cycle of power switch transistor Q1, the primary windingcurrent ramps up from zero to a peak winding current value that dependsupon the input voltage Vin, the switch on time, and the inductance forprimary winding 210. When power switch transistor Q1 cycles off, asecondary winding current flows across a second winding 225 intransformer 215, starting from a peak value and continuing to ramp downto zero. An output capacitor C2 and a resistor R2 stabilize a resultingoutput voltage Vout produced by the second winding current. A diode D1prevents the secondary winding current from flowing while the primarywinding conducts. Alternatively, diode D1 may be replaced by atransistor switch as is conventional for a flyback converter withsynchronous rectification. The secondary winding current produces areflected voltage across primary winding 210 and also across anauxiliary winding 230 for transformer 215. Diode D1 will have a voltagedrop across it that prevents a direct relationship between the reflectedvoltage and the output voltage while the secondary current still flows.But when the secondary current ramps to zero (the transformer resettime), there is no voltage drop across diode D1 such the resultingreflected voltage at that time is directly related to the outputvoltage. By controlling the reference voltage to track the reflectedvoltage across auxiliary winding 230 at this transformer reset time,controller 205 thus may determine the output voltage Vout as known inthe primary-only feedback arts as represented by the reference voltage.For example, controller 205 may include a Vsense pin 2 that samples thereflected voltage across auxiliary winding 230 through a voltage dividerformed by a pair of resistors R3 and R4. The reflected voltage may alsobe rectified through a diode D2 and a capacitor C3 to form a powersupply voltage VCC received by controller 205 at a power pin 1.

Controller 205 may include a ground pin 6 and a current sense (Isense)pin 4 that samples the primary winding current through the voltageproduced across a sampling resistor Rs coupled to a source of powerswitch transistor Q1. In some embodiments, controller 205 may determinethe rectified input voltage Vin indirectly through the Isense pinvoltage. However, controller 205 may also include a Vin pin 3 fordirectly sampling the rectified input voltage Vin through one or moreresistors R1.

Controller 205 may then compare the Isense voltage to the currentthreshold to determine whether the reflected voltage Vsense is trustableas shown in FIG. 3 for three consecutive cycles of the rectified inputvoltage Vin. From cycle to cycle of the rectified input voltage, thereis a dead period 301 during which the rectified input voltage amplitudeis relatively low. Outside of these dead periods, the rectified inputvoltage amplitude is relatively high such that relatively large amountsof power are pulsed to the load with each cycle of the power switch. Forexample, consider an initial period 300 for an initial cycle of therectified input voltage. The on-time periods 305 for two cycles of thepower switch (not illustrated) are shown in FIG. 3 for the initialperiod 300. Because of the relatively robust rectified input voltageduring the initial period 300, each on-time period 305 of the powerswitch produces a peak winding current as represented by the Isensevoltage that exceeds a current threshold 310. The correspondingreflected voltages as represented by Vsense pulses 315 from the on-timeperiods 305 will thus accurately reflect the output voltage at thetransformer reset time (TRST). Controller 205 (FIG. 2) may thus proceedto control the reference voltage in a conventional fashion duringperiods 300.

But current threshold 310 is not satisfied during dead periods 301. Forexample, consider an initial one of dead periods 301. The on-timeperiods 320 for the power switch during this initial dead period 301will not produce an Isense voltage that exceeds current threshold 310.The corresponding reflected pulses 325 thus no longer have an accuraterelationship to the output voltage at the transformer reset times as isthe case for trustable reflected pulses 315. But this problem is solvedby extrapolating Vref during the dead periods as discussed earlier.

A block diagram for the relevant portions of controller 205 for carryingout the adaptive extrapolation of the reference voltage during the deadperiods is shown in FIG. 4. Controller 205 includes a conventional Vreftracking circuit 401 that functions to adapt the reference voltage Vrefresponsive to the amplitude of the Vsense voltage at the transformerreset time. The detection of the transformer reset time may beaccomplished using a variety of techniques. For example, Vref may becompared to Vsense at a first comparator 410. Similarly, Vref may besummed with a positive offset voltage Δ (e.g., tens of millivolts) andthe resulting sum voltage compared with Vsense in a second comparator405. At the transformer reset time, a desired time gap will existbetween when second comparator 405 detects that the sum voltage and theVsense are equal as compared to when first comparator 410 detects thatthe reference voltage Vref and Vsense are equal. Should the referencevoltage Vref be too high, this time gap will increase above the desiredvalue. Conversely, if the reference voltage Vref is too low, the timegap will decrease below the desired value. A gap judgment module 415compares the current time gap between the assertions of the outputsignals from comparators 405 and 410 to the desired time gap value.Based upon the comparison in gap judgment module 415, a referencevoltage adjustment module 420 adjusts the amplitude of the referencevoltage Vref. For example, if the current gap value is too large,reference voltage adjustment module 420 may lower the amplitude for thereference voltage Vref. On the other hand, should the current gap valueequal the desired gap value (or be within a desired tolerance),reference voltage adjustment module 420 may leave the amplitude of thereference voltage unchanged. Finally, if the current gap value is toosmall as compared to the desired gap value, reference voltage adjustmentmodule 420 may increase the amplitude of the reference voltage Vref. Inthis fashion, the gap judgment causes the reference voltage Vref totrack the output voltage when Vsense is trustable. It will beappreciated that other techniques to track the reference voltage Vrefwith regard to the output voltage Vout may be used during the trustableperiods of Vsense.

A comparator 425 may be used to determine when Isense exceeds thecurrent threshold to assert an Isense good command 426 during thetrustable periods of Vsense. An extrapolator 430 samples the referencevoltage responsive to the de-assertion and assertion of Isense goodcommand 426 to determine the current first trustable Vref value andcurrent last trustable Vref value during each cycle of the rectifiedinput voltage. Based upon the current expired fraction of the deadperiod as discussed earlier and the current first and last trustableVref values, extrapolator 430 may generate a current extrapolated Vrefvalue 431 such as by using the linear or curvilinear extrapolationtechniques discussed earlier.

Controller 205 includes a primary-only feedback control circuit 435 thatoperates in a conventional fashion responsive to the reference voltageVref and Isense during the trustable periods in which Isense goodcommand 426 is asserted. The resulting primary only feedback control maybe based upon a peak current control or a constant on time control ofthe power switch as known in the power converter arts. The resultingswitch control may control the on time (ton), the off time (toff) or thepower switch period (tp) as is also known in the power converter arts.Feedback control circuit 435 continues to perform the same conventionalcontrol when Isense good command 426 is not asserted except that theextrapolated reference voltage 431 is substituted in place of thereference voltage. In this fashion, the resulting regulation is muchmore accurate during the non-trustable periods when Isense good command426 is not asserted.

Since the values of the current first and last trustable referencevoltages are used to extrapolate from the current last trustablereference voltage, it may be the case that the load demand changesduring the dead period extending from the current last trustablereference voltage. In other words, the past behavior of the load demandis used to predict the future behavior of the load demand. Such anassumption is relatively accurate for substantially static loads such asa powered LED. But it may be the case that the assumption introducessome inaccuracies due to vagaries in the load demand. For example,Figure SA illustrates the Vref extrapolation in the dead period betweenthe last trustable Vref value in an initial cycle of the rectified inputvoltage and a subsequent cycle of the rectified input voltage. In thesubsequent cycle, the load demand has increased such that a subsequentfirst trustable value 515 of Vref is higher than the minimum valuereached by extrapolated Vref 500. There is thus a sudden increase 505between the minimum value for extrapolated Vref 500 and subsequent firsttrustable Vref value 515. The converse situation is shown in FIG. 5B foran initial cycle and a subsequent cycle of the rectified input voltagein which the load demand decreases from the initial cycle to thesubsequent cycle. The minimum value during the dead period between thetwo cycles for an extrapolated Vref 520 will thus be too high ascompared to a subsequent first trustable value 530 of Vref. There willthus be a sudden decrease 535 from the minimum value for extrapolatedVref 520 to subsequent first trustable value 530 of Vref. Although thesechanges in the load demand produced undesirable inaccuracies in theextrapolated Vref, note that the extrapolation is adaptive. In otherwords, the subsequent first trustable value of Vref becomes the currentfirst trustable value of Vref for the next cycle of the rectified inputvoltage. The extrapolation will thus adapt to these load changes so asto accurately regulate the output power accordingly.

Although the preceding discussion was directed to a flyback convertertopology, note that it is conventional to indirectly sense the outputvoltage in other types of switching power converters using a reflectedvoltage on an auxiliary winding such as practiced in buck switchingpower converters or boost switching power converters. Such switchingpower converters would thus include a tracking circuit equivalent toVref tracking circuit 401 discussed above that would toggle a referencevoltage based upon the sense voltage from the auxiliary winding so thatthe reference voltage tracks the output voltage. These other switchingpower converter topologies would thus suffer the same problem ofnon-trustable “runt” pulses on the auxiliary winding during the deadperiods for the rectified input voltage. For example, a genericswitching power converter 600 is illustrated in FIG. 6 that may be abuck switching power converter, a boost switching power converter, abuck/boot switching power converter, or any other suitable switchingpower converter that uses a sense voltage derived from an auxiliarywinding 605. Controller 205 is configured to extrapolate the referencevoltage during the dead periods as discussed herein to control thecycling of a power switch 610. Accordingly, the reference voltageextrapolation techniques discussed herein are widely applicable to anysingle-stage switching power converter that tracks the output voltageindirectly using a sense voltage from an auxiliary winding.

As those of some skill in this art will by now appreciate and dependingon the particular application at hand, many modifications, substitutionsand variations can be made in and to the materials, apparatus,configurations and methods of use of the devices of the presentdisclosure without departing from the spirit and scope thereof. In lightof this, the scope of the present disclosure should not be limited tothat of the particular embodiments illustrated and described herein, asthey are merely by way of some examples thereof, but rather, should befully commensurate with that of the claims appended hereafter and theirfunctional equivalents.

We claim:
 1. A method of controlling a switching power converter,comprising: rectifying an AC mains voltage to produce a rectifiedsinusoidal voltage at the input of a primary winding of a transformerwhile cycling a power switch connected to the primary winding, whereincycling the power switch comprises cycling the power switch through aseries of power switch cycles, wherein each power switch cycle includesan on time for the power switch and an off time for the power switch;during the on time for each power switch cycle, determining whether acurrent through the power switch is greater than a threshold value toclassify each power switch cycle into either a trustable power switchcycle in which the current through the power switch is greater than thethreshold value or into a non-trustable power switch cycle in which thecurrent through the power switch cycle is not greater than the thresholdvalue; during the off time for each trustable power switch cycle,sensing a primary-only feedback voltage using primary-only feedback toregulate the cycling of the power switch; and during the off time foreach non-trustable power switch cycle, interpolating from theprimary-only feedback voltage to regulate the cycling of the powerswitch.
 2. The method of claim 1, wherein the non-trustable power switchcycles are organized into a series of dead periods, and wherein eachdead period follows a corresponding last one of the trustable powerswitch cycles, and wherein during the off time for each non-trustablepower switch cycle in each dead period, the interpolating from theprimary-only feedback voltage comprises interpolating from theprimary-only feedback voltage for the last trustable power switch cyclefor the dead period.
 3. The method of claim 2, wherein the interpolatingfrom the primary-only feedback voltage for the last trustable powerswitch cycle comprises linearly extrapolating from the primary-onlyfeedback voltage for the last trustable power switch cycle.
 4. Themethod of claim 2, the interpolating from the primary-only feedbackvoltage for the last trustable power switch cycle comprisescurvilinearly extrapolating from the primary-only feedback voltage forthe last trustable power switch cycle.
 5. The method of claim 1, whereinadjusting the reference voltage causes the reference voltage to track anoutput voltage of a flyback converter.
 6. A switching power converter,comprising: a comparator configured to compare a peak current through apower switch during an on-time for a power switch cycle of the powerswitch to a current threshold to classify the power switch cycle into aeither a trustable power switch cycle in which the peak current isgreater than the current threshold or into a non-trustable power switchcycle in which the peak current is less than the current threshold; aprimary-only feedback circuit configured to sense a primary-onlyfeedback voltage during an off time for each trustable power switchcycle; and an extrapolator configured to extrapolate from theprimary-only feedback voltage for a previous one of the trustable powerswitch cycles to produce an extrapolated voltage for a non-trustablepower switch cycle following the previous trustable power switch cycle.7. The switching power converter of claim 6, wherein the power switchcomprises a MOSFET.
 8. The switching power converter of claim 6, whereinthe power switch comprises a bipolar junction transistor.
 9. Theswitching power converter of claim 6, wherein the switching powerconverter is selected from a group consisting of a flyback converter, abuck converter, and a boost converter.
 10. The switching power converterof claim 6, further comprising a control circuit configured to controlthe cycling of the power switch during each non-trustable power switchresponsive to the extrapolated voltage.
 11. The switching powerconverter of claim 6, wherein the primary-only feedback circuit isconfigured to sense the primary-only feedback voltage from an auxiliarywinding for a transformer.
 12. The switching power converter of claim 6,wherein the extrapolator is further configured so that the extrapolationof the extrapolated voltage is a linear extrapolation.
 13. The switchingpower converter of claim 6, wherein the extrapolator is furtherconfigured so that the extrapolation is a piece-wise linearextrapolation.
 14. The switching power converter of claim 6, wherein theextrapolator is further configured so that the extrapolation is acurvilinear extrapolation.